1. Field of the Invention
The present invention relates to a voltage-frequency conversion apparatus and a method of changing a reference voltage of the voltage-frequency conversion apparatus, which are suitable for detecting the remaining voltage of electricity charged in a secondary battery, for example.
2. Description of the Related Art
Configuration of Conventional Voltage-Frequency Conversion Apparatus
<Overall Configuration>
With reference to FIG. 5, an example configuration of a conventional voltage-frequency conversion apparatus is described. FIG. 5 is a circuit block diagram showing the example configuration of a conventional voltage-frequency conversion apparatus.
A voltage-frequency conversion apparatus 100 shown in FIG. 5 has an error amplifier 102, a current source IA, variable current sources IB, IC, resistances RA, RB, p-type MOSFETs 104, 106, a reference voltage source 108, a comparator 110, a capacitor 112, a switch element 114 and control logic circuit 116.
The current source IA, the resistance RA and the source to drain of the p-type MOSFET are serially connected between the power supply VDD and ground. When the gate of the p-type MOSFET 104 is applied with a voltage VIN(−) with the resistance RA supplied with a current from the current source IA, a voltage V1 is produced at one end of the resistance RA. The value of this voltage level V1 is decided by the value of the voltage VIN(−). Whereas the variable current source IB, the resistance RB, p-type MOSFET 106 are serially connected between power supply VDD and ground. When the gate of the p-type MOSFET 106 is applied with a voltage VIN(+) with the resistance RB supplied with a current from the current source IB, a voltage V2 is produced at one end of the resistance RB. The value of this voltage V2 is decided by the value of the voltage VIN (+).
An input of the error amplifier 102 is connected to one end of the resistance RA at which the voltage V1 is produced, and the other input is connected to one end of the resistance RB at which the voltage V2 is produced. In other words, the error amplifier 102 performs a negative feedback operation to equalize voltages V2 and V1 according to the differential voltage between voltages V1, and V2. The variable current source IB is controlled by the output voltage from the error amplifier 102 to adjust the amount of current. Here, in the case the error amplifier 102 performs a negative feedback operation on the variable current source IB to equalize voltages V2 and V1, for example, when the voltage V2 becomes lower than voltage V1 and the differential voltage between voltages V1 and V2 increases due to the voltage VIN (+) becoming lower than the voltage VIN (−), the error amplifier 102 produces an output voltage to increase the amount of current of the variable current source IB so as to null the difference between voltages V1 and V2.
The amount of current generated from the variable current source IC is controlled according to the output voltage from the error amplifier 102. In other words, when the error amplifier 102 produces an output voltage to increase the current of the variable current source IB, the variable current source IC generates a larger current according to this output voltage. The variable current source IC and the capacitor 112 are serially connected between the power supply VDD and ground, and thus the capacitor 112 is charged by the current generated from the variable current source IC. In other words, the larger the current generated from the variable current source IC is, the quicker the capacitor 112 is charged whereas the smaller the current generated from the variable current source IC is, the slower the capacitor 112 is charged.
The comparator 110 compares a charging voltage occurring at one end of the capacitor 112 on the non-grounded side and a constant reference voltage VREF generated by the reference voltage source 108. In FIG. 5, the charging voltage of the capacitor 112 is applied to a plus (non-inverting input) terminal of the comparator 110 and the reference voltage VREF is applied to a minus (inverting input) terminal of the comparator 110. Therefore, the comparator 110 outputs a low level if the charging voltage of the capacitor 112 is smaller than the reference voltage VREF and outputs a high level if the charging voltage of the capacitor 112 exceeds the reference voltage VREF. In other words, the comparator 110 outputs a rectangular frequency signal corresponding to the differential voltage between the voltages V1 and V2.
The switch element 114 is connected in parallel with the capacitor 112. A bipolar transistor, a MOSFET, etc., can be employed as the switch element 114.
The control logic circuit 116 is connected to the output of the comparator 110 and controls the turning on/off of the switch element 114. In other words, the control logic circuit 116 renders the switch 114 on for a certain period after the output of the comparator 110 becomes a high level. In this period, the capacitor 113 is discharged through the switch element 114.
Operation of Conventional Voltage-Frequency Conversion Apparatus
With reference to FIG. 5 and FIG. 7, the operation of the voltage-frequency conversion apparatus 100 is described. FIG. 7 is a waveform diagram showing a relationship between the charging voltage appearing at one end of the capacitor 112 and a frequency signal output from the comparator 110. A rate (gradient) of increase in the charging voltage of the capacitor 112 varies according to the amount of the current supplied from the variable current source IC. In other words, as the current supplied from the variable current source IC becomes larger, the gradient of the increasing charging voltage of the capacitor 112 changes in a direction of becoming steeper, and the smaller the current supplied from the variable current source IC, the gradient becomes more gradual.
For example, an operation when the charging voltage takes a waveform in period TA in FIG. 7 is described. When there is a relationship that voltage VIN(+) equals VIN(−) +/− ΔV, in other words, when the differential voltage between voltages VIN(+) and VIN(−) is ΔV, a differential voltage of V1−V2 which corresponds to the differential voltage ΔV, is initially generated. Then, the error amplifier 102 produces an output voltage to null the differential voltage V1−V2. For example, when the differential voltage ΔV is positive, there is a relationship that voltage V1<voltage V2 and therefore, the error amplifier 102 produces an output voltage to reduce the current supplied from the variable current source IB. Also, this output voltage increases the current supplied from the variable current source IC. On the other hand, when the differential voltage ΔV is negative, there is a relationship that voltage V1>voltage V2 and therefore, the error amplifier 102 produces an output voltage to increase the current supplied from the variable current source IB. Also, this output voltage reduces the current supplied from the variable current source IC. In this way, the variable current source IC generates a current according to the output voltage supplied from the error amplifier 102, and the capacitor 112 is charged by the current supplied from the variable current source IC. Thus the charging voltage of the capacitor 112 increases with a gradient as in period TA to null the differential voltage between voltages V1 and V2. When the charging voltage of the capacitor 112 is lower than the reference voltage VREF, the output of the comparator 110 is at a low level.
Thereafter, when the charging voltage of the capacitor 112 exceeds the reference voltage VREF, the output of the comparator 110 becomes a high level. The control logic circuit 116 renders the switch element 114 on for a certain period after the output of the comparator 110 becomes the high level. In other words, a discharge path is formed for the capacitor 112. Therefore, the capacitor 112 immediately discharges through the switch element 114. The certain period in which the control logic circuit 116 renders the switch element 114 on is a period required for completing the discharge of the capacitor 112 and is decided in consideration of the capacity of the capacitor 112, etc. and preset in the control logic circuit 116. If the charging voltage of the capacitor 112 becomes smaller than the reference voltage VREF, the output of the comparator 110 becomes the low level again. Thus, the comparator 110 outputs to the capacitor 11 a frequency signal with a period of TO in response to the charging voltage.
Consequently, the voltage-frequency conversion apparatus 100 converts the differential voltage between voltages VIN(+) and VIN(−) into a frequency signal that corresponds to the differential voltage.
Example Application of Voltages V(+) and V(−)
The voltage-frequency conversion apparatus 100 can be employed as an apparatus for determining the remaining voltage of electricity charged in a secondary battery for example.
FIG. 6 is a schematic configuration diagram showing a battery pack incorporating a secondary battery. In FIG. 6, the battery pack 200 incorporates a secondary battery 201, a sensing resistor 202, a microcomputer 203 (or a logic integrated circuit), etc. The secondary battery 201 and the sensing resistor 202 are serially connected between a plus terminal and a minus terminal to be electrically connected to an electronic device using the secondary battery 201 as a power source. When the secondary battery 201 is charged or discharged, the sensing resistor 202 produces voltages VIN(+) and VIN(−) at both its ends. For example, if the battery pack 200 is mounted in an electronic device, the secondary battery 201 discharges to supply power to the electronic device and a discharge current flows in an a-direction (upwards on the page) through the sensing resistor 202. In other words, when the secondary battery 201 discharges, the voltage VIN(+) becomes lower than the voltage VIN(−). The smaller the discharging amount of the second battery 201 is, the greater the differential voltage between the voltage VIN(+) and the voltage VIN(−) becomes. On the other hand, if the battery pack 200 is mounted in a charger (not shown), the secondary battery 201 is charged and a charging current flows in a b-direction (downwards on the page) through the sensing resistor 202. In other words, when the secondary battery 201 is charged, the voltage VIN(+) becomes higher than the voltage VIN(−). The larger the charging amount of the second battery 201 is, the greater the differential voltage between the voltage VIN(+) and the voltage VIN(−) becomes.
The voltages VIN(+) and VIN(−) are supplied to the microcomputer 203 as voltage information which is the basis for determining the remaining voltage of electricity when the secondary battery 201 discharges or the charging voltage when the secondary battery 201 is charged. The microcomputer 203 incorporates the voltage-frequency conversion apparatus 100. Hence, the microcomputer 203 can detect the levels of the voltages V1 and V2 that are produced when voltages VIN(+) and VIN(−) are applied, as well as the differential voltage between voltages V1 and V2, and obtain a frequency signal according to the amount of current of the variable current source IC that nulls the differential voltage. The microcomputer 203 performs appropriate program processing on the obtained frequency signal thereby obtaining the remaining voltage of electricity and the usable time for the remaining voltage of electricity when the secondary battery 201 is mounted in an electronic device, and the charging voltage while being charged. (Refer to, e.g., Japanese Patent Application Laid-open Publication No. 2002-107428.)
The resistors RA and RB used in the voltage-frequency conversion apparatus 100 have a temperature characteristic that their resistances vary depending on the temperature of the resistors themselves or the ambient temperature.
When the differential voltage between voltages VIN(+) and VIN(−) is at ΔV, and the differential voltage V1−V2 between voltages V1 and V2 decided by voltages VIN(+) and VIN(−) is not null, the error amplifier 102 produces an output voltage to null the differential voltage V1−V2. The capacitor 112 is charged by a current provided from the variable current source IC. It is assumed that, for the case where the resistances of resistors RA and RB do not vary depending on the temperature, the increase gradient of the charging voltage of the capacitor 112 is, for example, the gradient in period TA. In other words, when the resistances RA and RB do not vary depending on the temperature, the comparator 110 is in a state of outputting a frequency signal with a period of TO.
When the resistance of at least either resistor RA or resistor RB varies depending on the temperature from this state, although the differential voltage between voltages VIN(+) and VIN(−) is ΔV, the output voltage of the error amplifier 102 shifts from an output voltage value that nulls the differential voltage V1−V2 corresponding to the differential voltage ΔV. As a result, the problem occurs that the frequency of the frequency signal output from the comparator 110 deviates from the true frequency.
As an illustrative example, in period TB of FIG. 7, a case when the resistance of resistor RB increases depending on its own temperature (e.g. when the resistance value of resistance RB doubles) is considered. When the resistance of resistor RB doubles due to the temperature characteristic, a change is made from a state where voltage V2 is equal to voltage V1, which is achieved by the negative feedback operation by the error amplifier 102 that produces an output voltage to null the differential voltage between voltages V1 and V2, to a state where voltage V2 is higher than voltage V1. Accordingly, the error amplifier 102 produces an output voltage to change the current supplied from the variable current source IB to a current value that is half the previous value in order to bring back voltage V2 to become equal to voltage V1. Hence, not only the current value of the current supplied from the variable current source IB but also that from the variable current source IC is changed to half its previous value. As a result, since the current per unit of time supplied from the variable current source IC to the capacitor 112 is reduced to half, the rise gradient while the capacitor 112 is being charged is half of that in period TA. Therefore, although the differential voltage between voltages VIN(+) and VIN(−) is ΔV, the problem occurs of faulty generation of a frequency signal from the comparator 110 with a period of 2TO that is twice the true period of TO (dashed line).
For example, when the voltage-frequency conversion apparatus 100 is incorporated in a microcomputer 203 used in the battery pack 200 and the battery pack 200 is mounted in an electric device, the remaining voltage of electricity in the secondary battery 201 can be detected. However, since the microcomputer 203 determines the remaining voltage of electricity based on the frequency of the frequency signal obtained from the comparator 110, if such a false frequency signal is generated, the determined remaining voltage of electricity is quite apart from a true remaining voltage of electricity and thus causes a great deal of trouble for the user of the electronic device.
FIG. 8 is a graph showing an input-output characteristic of a voltage-frequency conversion apparatus. In FIG. 8, the horizontal axis shows a differential voltage [V] between the voltages VIN(+) and VIN(−), and the vertical axis shows the frequency signal [Hz] output from the comparator 110. The differential voltage between the voltages VIN(+) and VIN(−) and the frequency signal ideally has proportionality as shown by the solid line. However, when at least either resistor RA or resistor RB varies depending on the temperature, the relationship of the differential voltage between the voltages VIN(+) and VIN(−) with the frequency of the frequency signal output from the comparator 110 deviates from the characteristic shown by the solid line to the dot-and-dash line side or the dashed line side.